The invention relates to amplifiers, especially monolithic integrated circuit amplifiers, and more particularly to such amplifiers having very high gain-bandwidth products, and more particularly to instrumentation amplifiers having very high gain bandwidth products.
A "classical" single pole operational amplifier has a gain given by the equation: ##EQU1##
In this equation, V.sub.in is the "small signal" or AC input voltage, and V.sub.out is the AC output voltage. C.sub.s is the capacitance of the compensation capacitor, and g.sub.m is the transconductance of the amplifier. Those skilled in the art know that the gain-bandwidth product for this type of circuit is essentially constant. This means that as the closed loop gain of the amplifier is increased, its bandwidth decreases proportionately, making it very difficult to obtain high gain at high frequencies.
Instrumentation amplifiers are commonly used to amplify small differential input signals to produce differential or single ended output signals. Such instrumentation amplifiers commonly utilize two single ended amplifiers coupled together by a common gain-setting resistor plus a differential-to-single ended converter or a difference amplifier.
A significant recent advance in improving the gain-bandwidth product of amplifiers and instrumentation amplifiers is thoroughly described in "A Programmable Instrumentation Amplifier for 12-Bit Resolution Systems", by Wurcer and Counts, pages 1102-1111 of the IEEE Journal of Solid State Circuits, Vol. SC-17 No. 6, December 1982, incorporated herein by reference. FIG. 1 in that article shows a simplified instrumentation amplifier, and FIG. 2 shows a complete schematic of an instrumentation amplifier. The AC performance of the individual amplifiers which form the input stages of the instrumentation amplifier is described in detail on pages 1109 and 1110. An improved feedback amplifier circuit having a greatly improved gain-bandwidth product only at low values of closed loop gain is disclosed and described. In the described circuit, the input signal is applied to the base of an NPN transistor, the emitter of which is connected to a summing node. A gain-setting resistor R.sub.G is connected between the summing node and ground. The collector of the NPN transistor is coupled to a resistive load (such as a PNP current source), and also to the negative input of an operational amplifier having gain A. The output of the operational amplifier is coupld back to the negative input of the operational amplifier by a capacitor C and is connected by a resistor R.sub.F back to the summing node. The effective g.sub.m for this circuit is given by the equation ##EQU2##
It can be shown that this expression can be re-written as ##EQU3## where A.sub.CL is the closed loop gain of the amplifier. r.sub.e is the dynamic emitter resistance of the NPN input transistor, and typically has a value of 500 ohms. The feedback resistance R.sub.F typically has a value of about 20 kilohms. Consequently, it can be seen that for low values of closed loop gain A.sub.CL, below approximately 40, r.sub.e is negligible, and therefore the effective gain of the improved feedback amplifier is given by the equation ##EQU4## However, for high values of A.sub.CL, the r.sub.e term dominates, and then the effective transconductance is given by the expression ##EQU5##
Thus, it can be seen that for low closed loop gains, the transconductance of the amplifier is not constant, as in the "classical" case, but instead is proportional to the closed loop gain of the amplifier. As the gain is increased, the bandwidth is also increased, and the gain-bandwidth product is also increased. The unity gain frequency is also increased. Those skilled in the art will recognize that what this really means is that the bandwidth stays constant as the closed loop gain is increased, as long as the r.sub.e term is negligible. However, when the closed loop current gain A.sub.CL increases beyond approximately 40, so that the r.sub.e term is no longer negligible, from that point on, the circuit behaves like the "classical" circuit in that the gain-bandwidth product is constant. Further increases in the gain A.sub.CL are made at the expense of bandwidth, and at very high frequencies very little amplifier gain is available compared to amounts available at low frequencies.
Nevertheless, the circuit described in the above-mentioned Wurcer-Counts paper has been commercially very successful, and in many of its applications there has been no need to sacrifice gain for bandwidth. When increased gain has been needed, it has been possible to extend the range of A.sub.CL in which equation (4) above holds by increasing the emitter current of the NPN transistor, thereby reducing r.sub.e. However, there is a definite limit to which this is practical, because increasing the emitter currents increases the power dissipation and increases the base currents of the input transistors, reducing the input impedance of the amplifier, and, especially when transducers having high output impedance are connected to the base of the input transistors, results in amplification of noise components of the base currents. Furthermore, mismatches in the increased base currents would result in increased offset voltages between the amplifier inputs.
Thus, there remain numerous applications in which it would be desirable to have a feedback amplifier that avoids the above-described "classical" gain-bandwidth trade-off problem, not only for low closed loop gains, but also for high closed loop gains.